Power supply for microwave discharge light source

ABSTRACT

A power supply circuit for a magnetron adapted to supply microwave energy to an electrodeless discharge bulb is disclosed. The circuit comprises a rectifier coupled across a commercial AC voltage source, a filter for smoothing the output of the rectifier, an inverter for converting the DC voltage supplied from the filter into a high frequency AC voltage, a step-up transformer for stepping up the high frequency AC voltage outputted from the inverter, and a rectifier which rectifies the high voltage AC output of the transformer into a unidirectional voltage which is supplied to the magnetron. The inverter switching is controlled by a pulse width modulation control circuit to maintain the magnetron output power at a predetermined level. According to one aspect, an inductance is provided in the circuit which supresses high frequency components in the currents flowing through the windings of the transformer; according to another aspect, the inverter switching frequency (expressed in kHz) is set at a value not less than 1500/D, wherein D represents the diameter of the electrodeless bulb expressed in millimeters; according to still another aspect, the peak to the mean value ratio of the magnetron current is limited under 3.75 inclusive.

TECHNICAL FIELD

The present invention relates to a microwave generating system includinga magnetron and a power supply circuit therefor, which is adapted tosupply microwave energy to a microwave discharge light source, includingan electrodeless bulb.

BACKGROUND ART

In recent years, microwave discharge light source having anelectrodeless bulb disposed in a microwave resonance cavity has beendeveloped and is attracting attention because of its long life. FIG. 1ashows one of such microwave discharge light source apparatus disclosedin Japanese Laid-Open Patent Application No. 56-126250; FIG. 1b shows amodification thereof disclosed in Japanese Laid-Open Patent ApplicationNo. 57-55091. In both apparatuses, a magnetron 1 having an antenna 1a isdisposed at the end of a waveguide 2 having ventilating holes 2a whichsupplies the microwave generated by the magnetron 1 to a resonancecavity 3 through a microwave supply port 3a; the cavity 3 is formed by aparaboloidal wall 3b having a light reflecting rotationally symmetricinner surface and a metallic mesh 3c forming the front face of thecavity 3, which opaque to microwave but transparent to light. Aspherical electrodeless discharge bulb 4 disposed in the cavity 3 andhaving encapsulated therein a plasma generating medium emitts lightthrough the metallic mesh 3c covering the front face of the cavity 3,when the microwave is radiated into the bulb 4: at first, the gasenclosed in the bulb 4 undergoes discharge due to the microwave radiatedinto the cavity 3; thus, the inner surface of the bulb 4 is heated, andthe metal, such as mercury, deposited on the inner surface of the bulb 4is evaporated into a gas; as a result, the discharge in the bulb 4 goesover to that of the metallic gas, in which light having an emissionspectrum peculiar to the kind of the metal is emitted from thedischarging metallic gas. The emitted light is reflected by the cavitywall 3b and is radiated forward through the front mesh 3c. Theapparatuses further comprise a fan 5 at the end wall of the housing 6for cooling the magnetron 1 and the bulb 4.

Microwave discharge light source apparatuses similar to those describedabove are also disclosed in U.S. Pat. Nos. 4,498,029 and 4,673,846, bothissued to Yoshizawa et al. The first of these U.S. Patents teach anapparatus in which the bulb is sufficiently small to act substantiallyas a point light source; the second teach an apparatus in which the wallsurface of the microwave resonance cavity having the electrodeless bulbdisposed therein is mostly constituted by a mesh, wherein the wiresconstituting the mesh are electrically connected each other without anycontact resistance.

A conventional power supply circuit for a magnetron is disclosed inJapanese Laid-Open Utility Model Application No. 56-162899, or in thefirst of the above mentioned U.S. Patents, according to which acommercial voltage source at 50 to 60 Hz is coupled to a step-uptransformer, and the resulting stepped-up high-voltage AC current isrectified by a full-wave rectifier circuit to obtain pulsingunidirectional current which is supplied to the magnetron. As therectification is effected by a full-wave rectifier circuit, theresulting high voltage rectified current pulsates at 100 to 120 Hz;consequently, the magnetron generates a microwave pulsing at 100 to 120Hz. Thus, when magnetron 1 is supplied by this conventional circuit, thedischarge in the bulb 4 is caused by the microwave pulsing at 100 to 120Hz.

The disadvantage of this type of conventional power supply circuit is asfollows. First, as the commercial AC voltage of relatively lowfrequency, i.e., 50 to 60 Hz, is directly supplied to the primarywinding of the step-up transformer to obtain a high voltage needed tosupply the magnetron, the transformer should be provided with a heavyiron core; the weight of the transformer is equal to or greater than 10kg when the input power to the magnetron is 1.5 kW. Second, as afull-wave rectifier circuit is used to rectify the AC current induced inthe secondary winding of the transformer, neither one of the terminalsof the secondary winding can be grounded; thus, the over-all size of thetransformer should be further increased to ensure an electricalinsulation thereof; in addition, extremely high voltage may develop inportions within or outside of the transformer, which diminishes thereliability of the parts thereof. If the rectifier circuit coupled tothe secondary winding of the transformer is constituted by a half-waverectifier circuit, one terminal of the secondary winding of the step-uptransformer can be grounded to minimize the above-mentioned drawbacks ofthe conventional power supply circuit. This, however, causes anotherproblem: as the voltage applied to the magnetron 1 is reduced to 0during the half period of the commercial AC voltage cycle, thegeneration of the microwave is stopped for about 8 to 10 ms; thus thereis the danger that the discharge is extinguished during the same timeintervals. Thus, a full-wave rectifier circuit must have been used torectify the outputs of the step-up transformer.

FIG. 2a shows an inverter type power supply circuit for a magnetrontaught in Japanese Patent Publication No. 60-189889, wherein themagnetron 1 is supplied by the circuit as described in what follows. Arectifier circuit 8 is coupled across the lines of a commercial ACvoltage source E; a pair of series-connected capacitors C1 and C2 arecoupled across the output terminals of the rectifier circuit 8 to obtaina substantially constant voltage DC power. An oscillator circuit 9,which comprises a Zener diode Zn, a capacitor C3, a plurality ofresistors, and an amplifier A, is coupled across the capacitor C2 tooutput a rectangular waveform signal having a frequency substantiallyhigher than that of the commercial AC voltage source E to a controlcircuit 10 comprising a transistor T1, a diode D1, and a plurality ofresistors; the frequency of the rectangular waveform signal of theoscillator circuit 9 is determined by the values of the resistors andthe capacitor C3 thereof. The control circuit 10 controls the alternateswitching actions of a switching circuit comprising the powertransistors 11 and 12 and the controlling transistors 11a and 12atherefor. Namely, by alternately turning on and off the controllingtransistors 11a and 12a, the circuit 10 alternately turns on and off thepower transistors 11 and 12 in response to the output signal of theoscillator circuit 9. Thus, a high frequency rectangular waveform ACcurrent is supplied to the primary winding P of the transformer Tthrough a filter circuit 13. The AC voltage induced in the secondarywinding S of the transformer T is rectified by a voltage doublerrectifier circuit consisting of a capacitor C4 and a diode D2, and issupplied therefrom to the magnetron 1.

The inverter type power supply for a magnetron as described above alsosuffers disadvantages. Namely, as the magnetron 1 constitutes anon-linear load, the output power and current thereof and the invertercurrent supplied to the step-up transformer become unstable when thevoltage level of the voltage source E fluctuates; the over-currentresulting therefrom may destroy the power transistors 11 and 12.

FIG. 2b shows another inverter type power supply circuit for a magnetrontaught in Japanese Laid-Open Patent Application No. 62-113395, whereinthe magnetron 1 is supplied by the circuit as follows. A diode bridgerectifier circuit 8 comprising four diodes Do is coupled across thecommercial AC voltage source E; a smoothing filter circuit 9 consistingof a capacitor Co is coupled across the output terminals of therectifier circuit 8 to output a substantially constant DC voltagetherefrom. The switching circuit 10 comprises switching transistors Q1and Q2 and diodes D1 and D2 for reverse currents coupled across thesource and the drain thereof, respectively, the transistors Q1 and Q2being coupled across the negative output terminal of the filter circuit9 and the terminals P1 and P2 of the primary winding P of thetransformer T, respectively. The positive output terminal of the filtercircuit 9 is coupled to the center tap 0 of the primary winding P of thetransformer T. The gate terminals g1 and g2 of the transistors Q1 andQ2, respectively, is coupled to the center tap 0 of the primary windingP of the transformer T. The gate terminals g1 and g2 of the transistorsQ1 and Q2, respectively, are coupled to the output terminals of acontrol circuit 11. The voltage doubler rectifier circuit 12 consistingof series-connected capacitor C1 and a diode D3 is coupled across theterminals S1 and S2 of the secondary winding S of the transformer T; thenegative output terminal d of the rectifier circuit 12 is coupled to thecathode K of the magnetron 1, which is heated by a filament currentsupplied thereto from a commercial AC voltage source through anelectrically insulating transformer (not shown) and the lines h; thepositive output terminal f of the rectifier circuit 12, on the otherhand, is coupled to the anode A of the magnetron 1 through a resistor R,the terminals of the resistor R being coupled to the input terminals ofthe control circuit 11.

The control circuit 11 outputs pulses to the transistors Q1 and Q2 at avarying frequency centered around a fixed frequency, to alternately turnon and off the transistors Q1 and Q2. Thus, the current flowsalternately from the center tap 0 to the terminal P1 and to the terminalP2 of the primary winding P of the transformer T to induce an AC voltagein the secondary winding S thereof, which is rectified by the rectifiercircuit 12 and supplied therefrom to the magnetron 1. The pulse signalsof the control circuit 11 at the fixed frequency are subjected tofrequency modulation utilizing a modulating signal having a frequencywhich is lower than the frequency of the fixed frequency of the outputpulse signals, to prevent flickering of the discharge in anelectrodeless bulb such as those shown in FIGS. 1a and 1b; theflickering of the discharge is caused by an acoustic resonance in thebulb due to the ripple or fluctuation of the microwave energy. Further,the circuit 11 varies the length of time during which the transistors Q1and Q2 are turned on, so that the output power of the magnetron is heldconstant irrespective of the fluctuation in the voltage source level;this can be effected by detecting the magnetron current by means of thevoltage drop across the resistor R, thanks to the substantially constantvoltage characteristic of the magnetron 1.

The inverter type power supply circuit for a magnetron described justabove is small-sized and is effective to a certain degree to prevent theflickering of the discharge arc of the electrodeless discharge bulb,thanks to the adoption of the high frequency inverter in the circuit.The flickering of the discharge arc, however, may persist even in theapparatuses supplied by the circuit, depending on the kind and amount ofthe material encapsulated in the bulb and on the microwave energy levelradiated into the bulb: the flickering of the arc is particularlymanifest when a metal halide compound such as sodium iodide isencapsulated in the bulb in addition to mercury and a starter rare gas,or when the microwave energy supplied to the bulb is at a high level.Further disadvantage of the circuit of FIG. 2b is that the controllingcircuit 11 thereof has a complicated structure, because the pulsesignals thereof are subjected to frequency modulation and the length ofthe turning-on time of the switching is varied to maintain the outputpower of the mangetron 1 at a constant level.

Power supply circuits for a magnetron utilizing inverters are alsodisclosed in U.S. Pat. No. 4,593,167 issued to Nilssen and U.S. Pat. No.3,973,165 issued to Hester. The first of these U.S. patents teach apower supply circuit for a magnetron of a microwave oven including aninverter, wherein the step-up transformer exhibits relatively highleakage between its input and output windings and a capacitor isconnected across the step-up transformer's output winding; further, arectifier and filter means is connected in parallel with the capacitor,and supplies substantially constant DC voltage to the magnetron. Thesecond U.S. patent teach an inclusion of an inverter in a power supplyfor a magnetron which supplies microwave energy to a microwave oven,etc, wherein the DC current obtained by rectifying a commercial ACvoltage of 60 Hz is supplied to the step-up transformer through aninductor, which prevents high frequency currents or voltages to flowinto the AC voltage source lines. Further, Japanese Laid-Open PatentApplication No. 62-290098 teaches a microwave discharge light sourceapparatus including an inverter type power supply circuit for themagnetron, wherein the inverter frequency is set at a few tens kHz, forexample, thereby maintaining parameters of the plasma in the bulb at asubstantially constant level to prevent the flickering of the dischargein the bulb.

DISCLOSURE OF THE INVENTION

Thus, an object of the present invention is to provide a power supplycircuit including a magnetron adapted to supply microwave energy to amicrowave discharge light source apparatus including an electrodelessdischarge bulb, wherein the circuit is small in size and light inweight; more particularly, an object of the present invention is toreduce the size and weight of the step-up transformer comprised in thecircuit.

Another object of the present invention is to provide such power supplycircuit including a magnetron which supplies microwave energy that iscapable of sustaining stable discharge in the electrodeless bulb of thelight source apparatus; namely, it is an object of the present inventionto provide a power supply circuit which does not cause flickering in thedischarge in the bulb and which is capable of sustaining the dischargein the bulb without any fear of extinguishment.

According to the present invention, a power supply circuit systemincluding a magnetron adapted to supply microwave energy to a microwavedischarge light source apparatus including an electrodeless dischargebulb is provided, which comprises: rectifier and filter means, adaptedto to be coupled to a commercial AC voltage source, for supplying asubstantially constant DC voltage; inverter means, supplied by therectifier and filter means, for converting the DC voltage into a highfrequency AC voltage having a waveform of alternating pulses; pulsewidth modulation means for modulating the pulse width of the pulses ofthe AC voltage outputted by the inverter means; step-up transformerhaving an input or primary winding supplied by the output of theinverter means, the output or secondary winding thereof outputting astepped-up high frequency AC voltage, the voltage level of which issubstantially higher than that of the commercial voltage source; secondrectifier means, coupled to the secondary winding of the step-uptransformer, for rectifying the output voltage of the secondary windingof the step-up transformer into a DC voltage; and a magnetron suppliedwith the voltage outputted by the second rectifier means.

According to one aspect of the present invention, the circuit systemfurther comprises inductance means operatively coupled to the step-uptransformer to suppress the rapid changes in the level of the currentflowing through the primary or the secondary winding of the step-uptransformer. In other words, inductance means is provided which reduceshigh frequency components in the current flowing through the primary orthe secondary winding of the step-up transformer. Thus, stable operationof the inverter is ensured.

According to a second aspect of the present invention, the inverterswitching frequency, i.e., the frequency of the AC voltage outputtedtherefrom, expressed in Kiloherz is set at a value which is not lessthan 1500/D, wherein D is the diameter, expressed in millimeters, of theelectrodeless discharge bulb supplied by the magnetron of the circuitsystem. Thus, a stable discharge without flickering can be maintained inthe electrodeless bulb without any fear of extinguishment.

According to a third aspect of the present invention, the circuit systemfurther comprises high frequency component reducing means for reducingthe high frequency components of the magnetron current, thereby limitingthe ratio i_(max) /i_(o) of the peak to the means value of the magnetroncurrent under 3.75 inclusive:

    i.sub.max /i.sub.o <3.75

Thus, the flickering in the discharge can be effectively suppressed.

BRIEF DESCRIPTION OF THE DRAWINGS

Further details of the invention will become more clear in the followingdescription of the best modes for carrying out the present invention,taken in conjunction wit the accompanying drawings, in which:

FIGS. 1a and 1b are schematic sectional views of conventional microwavedischarge light source apparatuses;

FIGS. 2a and 2b are diagrams showing conventional power supply circuitsfor a magnetron, which may be installed to supply microwave energy to anapparatus shown in FIG. 1a or 1b;

FIG. 3a is a diagram showing a power supply circuit according to a firstembodiment of the present invention;

FIG. 3b is a block diagram showing the details of the PWM controlcircuit in the power supply circuit of FIG. 3a;

FIG. 4 shows waveform of voltages and currents in the circuit of FIG.3a;

FIG. 5 shows the curent-voltage characteristic of a magnetron

FIG. 6 shows the relationships between the pulse width of magnitudecorresponding to the output power of the mangetron;

FIG. 7 shows the relationships between the pulse width of the gatesignals supplied to the inverter switching circuit and a magnitudecorresponding to the peak magnetron current;

FIGS. 8 and 9 are diagrams showing power supply circuits for a magnetronaccording to the second and the third embodiment, respectively, of thepresent invention;

FIG. 10 is a diagram showing a power supply circuit for a magnetronaccording to the fourth embodiment of the present invention;

FIG. 11 shows a waveforms of the magnetron output power in the circuitof FIG. 10;

FIG. 12 is a diagram showing a power supply circuit for a magnetronaccording to a fifth embodiment of the present invention;

FIG. 13 shows waveforms of currents and voltages in the circuit of FIG.12;

FIG. 14 shows waveforms of mangetron currents in the circuit of FIG. 12;

FIG. 15 shows the relationship between the peak to the mean value ratioof the mangetron current and the intensity of flickering observed in thedischarge in the electrodeless discharge bulb; and

FIG. 16 shows the relationships between the inverter switching frequencyand the capacitance coupled across the magnetron which is effective insupressing the occurrence of flickering in the discharge in theelectrodeless bulb.

BEST MODES FOR CARRYING OUT THE INVENTION First Mode: FundamentalStructure and Operation

Referring now to FIGS. 3a and 3b of the drawings, a first embodimentaccording to the present invention is described.

The power supply circuit for the magnetron 1 comprises a diode bridgefull-wave rectifier circuit 2, the input terminals of which are coupledacross a commercially available AC voltage source E, typically on theorder of 100 to 220 volts RMS at 50 to 60 Hz. A voltage dividerconsisting of a pair of resistors R1 and R2 connected in series iscoupled across the output terminals of the rectifier circuit 2. Further,a capacitor C1 constituting a smoothing filter circuit is coupled acrossthe output terminals of the rectifier circuit 2 to supply asubstantially constant DC voltage therefrom. The input terminals of theinverter switching circuit comprising four MOSFETs (metal oxidesemiconductor field effect transistors) Q1 through Q4 connected inbridge circuit relationship are coupled across the output terminals ofthe filter circuit, the capacitor Cl; the output terminals of theswitching circuit is coupled across the primary or input winding P ofthe step-up transformer T having a step-up ratio of 1 to n, a reactor Lbeing inserted in series with the primary winding P. The inverterswitching circuit further comprises four diodes D1 through D4 forreverse currents, which are coupled across the source and the drainterminal of the MOSFETs Q1 through Q4, respectively, the gate terminalsof the MOSFETs being coupled to the output terminals of the PWM (pulsewidth modulation) control circuit 3. Further, a voltage doublerhalf-wave rectifier circuit consisting of a capacitor C2 and a diode D5connected in series is coupled across the secondary or output winding Sof the transformer T; the output terminals of the rectifier circuit,i.e., the terminals across the diode D5, are coupled across the cathodeK and the anode An of the magnetron 1 to supply a pulsating DC currenti_(Mg) thereto.

The output terminals of a current detector 4 for detecting the currentflowing through the secondary winding S of the transformer T are coupledto the PWM control circuit 3 to output a voltage Vf corresponding to thecurrent flowing through the secondary winding S. As, shown in FIG. 3b,the control circuit 3 comprises a half-wave rectifier 3a rectifying theoutput Vf of the current detector 4, a smoothing filter 3b coupled tothe output of the rectifier 3a to output a smoothed voltage Vf'corresponding to the mean value of the voltage Vf; the error detector orsubtractor 3d is coupled to the outputs of the filter 3b and a variableresistor 3c outputting a pre-set reference voltage Vr, and outputs thedifference:

    Ve=Vr-Vf'

between the reference Vr and the mean voltage Vf'. The amplifier 3eamplifies the error or the difference Ve by a factor A, and outputs an,amplified error signal:

    Ve'=A·Ve.

Further, for the purpose of feeding the value of the voltage Vo forwardto the control circuit 3, the output terminal of the voltage deviderconsisting of the resistors R1 and R2 i.e., the terminal at theintermediate position between the two resistors R1 and R2, which outputsa voltage Vin corresponding to the output voltage Vo of the smoothingfilter capacitor C1, is coupled to another amplifier 3g which amplifiesthe signal Vin by a factor of B to output a signal:

    Vb=B·Vin

The subtractor 3f coupled to the outputs of the amplifiers 3e and 3goutputs the difference

    Vp=Ve'-Vb

to the modulator 3h. The modulator 3h outputs pulses Vw at apredetermined fixed frequency which is substantially higher than that ofthe AC voltage source E, the width of the pulses Vw being modulated,i.e., varied with respect to a predetermined fixed pulse width, inproportion to the value of the signal Vp. The driver circuit 3i coupledto the output of the modulator 3h outputs gate signals to the MOSFETsQ1, through Q4 of the inverter switching circuit in response to thesignal Vw, and alternately turns on and off the MOSFETs Q1 and Q4 andthe MOSFETs Q2 and Q3. Thus, high frequency AC current flows through theprimary winding P of the transformer T to induce an AC voltage in thesecondary winding S thereof, which is rectified and supplied to themagnetron 1 through the rectifier circuit consisting of the capacitor C2and the diode D5.

More explicit description of the operation of the circuit of FIGS. 3aand 3b is as follows.

First, the operation during a positive half-cycle Tp of the inverterswitching cycle is described, referring to FIG. 4 as well as FIGS. 3aand 3b. When the driver 3i of the control circuit 3 turns on the MOSFETsQ1 and Q4, while the MOSFETs Q2 and Q3 are turned off, the outputvoltage V1 of the inverter switching circuit rises substantially to alevel equal to the output voltage Vo of the filtering capacitor C1 andis kept thereat during the time interval in which the MOSFETs Q1 and Q4are tuned on; thus, the output voltage V1 of the inverter switchingcircuit has a square-shaped waveform, as shown in FIG. 4(a). Theduration T_(ON) of the positive voltage V1 i.e., the pulse width thereofcorresponds to the pulse width of the gate signal outputted from thedriver 3i and that of the signal Vw outputted from the PWM modulator 3hof the control circuit 3; the height of the pulse V1 is substantiallyequal to the output voltage Vo of the filtering capacitor C1. Due to theinductance of the reactor L connected in series with the primary windingP of the transformer T, the current i₁ flowing through the primarywinding P in the direction shown by the arrow in FIG. 3a increasesgradually from zero to a maximum during the time in which the voltage V1is maintained at the positive level, as shown in FIG. 4(b); after theMOSFETs Q1 and Q4 are turned off and the voltage V1 returns to zerolevel, the current i₁ in the primary winding P of the transformerpersists during a short time Tx, due to the existance of the inductanceof the reactor L connected in series with the primary winding P. Duringthis short time period Tx, the current i₁ flows through the diodes D2and D3 to charge the capacitor C1. The current induced in the secondarywinding S of the transformer during this positive half-cycle Tp of theinverter has a polarity corresponding to the conducting direction of thediode D5; thus, no currents i_(Mg) flows through the magnetron 1 and thevoltage V2 across the cathode K and the anode An of the magnetron 1 isequal to zero, as shown in FIG. 4 (c) and (d), the capacitor C2 beingcharged by the current induced in the secondary winding S during thepositive half-cycle Tp.

The operation of the power supply circuit during the negative half-cycleTn of the inverter is as follows. During the negative half-cycle Tn, theMOSFETs Q2 and Q3 are turned on by the control circuit 3; thus, thepolarities of the output voltage V1 of the inverter switching circuitand the current i₁ flowing through the primary winding P of thetransformer T are reversed, as shown in FIGS. 4 (a) and (b). Except forthis, the operation of the circuit electrically coupled to the primarywinding P of the transformer T during the negative half cycle Tn issimilar to the operation thereof in the positive half-cycle Tp. However,the voltage induced in the secondary winding S by the current i₁ flowingthrough the primary winding P in the direction opposite to that shown bythe arrow in FIG. 3a, the induced voltage in the secondary winding S issuperposed on the voltage developed across the capacitor C2 which isalready charged in the preceding positive half-cycle Tp; thus, as shownin FIG. 4(c), the voltage V2 applied across the magnetron 1 jumps to thevoltage level to which the capacitor C2 has been charged in the previoushalf-cycle Tp, when the MOSFETs Q2 and Q3 are turned on and the outputvoltage V1 goes down from zero to a negative level as shown in FIG.4(a). After this, the voltage V2 applied across the mangetron 1increases gradually during the time T_(ON) in which the MOSFETs Q2 andQ3 are turned on and the output voltage V1 of the switching circuit iskept at the negative level, due to the gradual decrease of the voltagedeveloped across the reactor L during the same time period T_(ON). Thecurrent i_(Mg) flowing through the magnetron 1, on the other hand,increases gradually from Zero to a maximum, as shown in FIG. 4(d) duringthe time T_(ON), due to the current-voltage characteristic of themagnetron 1. Namely, as shown in FIG. 5, the voltage V2 across themagnetron 1 plotted along the ordinate is at a finite voltage level Vzwhen the magnetron current i_(Mg) plotted along the abscissa begins toflow through the magnetron 1. The magnetron voltage V2 increaseslinearly from this cut-off voltage Vz to a maximum Vz+ΔVz, as themagnetron current i_(Mg) increases from zero to i_(R), exhibiting theequivalent series resistance

    r.sub.Mg =ΔVz/i.sub.R

in the linear relationship range. After the MOSFETs Q2 and Q3 are turnedoff and the output voltage V1 of the inverter switching circuit returnsto zero level, the current i₁ in the primary winding P of thetransformer T persists in the short length of time Tx due to the reactorL, during which the magnetron voltage V2 and the magnetron currenti_(Mg) decreases and returns to the zero level at the end thereof, asshown in FIGS. 4 (c) and (d).

The output power of the magnetron 1 is held at a constant level by themodulation of the pulse width T_(ON) of the gate signals applied to theMOSFETs Q1 through Q4 from the control circuit 3. Detailed explanationthereof is as follows.

The output power P_(OUT) of the magnetron 1 is approximately given bythe product of the mean value of the magnetron current i_(Mg) shown inFIG. 4(d) and the magnetron voltage V2, because the rise ΔVz in thevoltage V2 is small compared to the magnitude of the cut-off voltage Vz,as shown in FIG. 5, when the magnetron 1 is operated within the ratedcurrent and voltage range. Thus, P_(OUT) is approximated as follows:##EQU1## wherein, the meanings of the symbols are as follows:

f: the switching frequency of the inverter, or the frequency of thepulses of the voltage V2 and the current i_(Mg) ; ##EQU2##

Ro: the interior resistance of the voltage source;

n: step-up ratio of the transformer T;

L: inductance of the reactor L;

C: the conversion value of the capacitance of the capacitor C4 in aequivalent circuit in which the capacitor C4 is forming part of thecircuit electrically coupled to the primary winding P;

T_(ON) : the length of time during which the MOSFETs Q1 through Q4 areturned on, which is equal to the pulse width of the output signals ofthe control circuit 3, or the pulse width of the voltage V1, as shown inFIG. 4(a);

the values of a and b in the equation (1) being given as follows:##EQU3##

Thus, FIG. 6 shows the relationship between the value ##EQU4## appearingin the right hand side of equation (1) and T_(ON), in the case where

n=10,

C=0.47×10⁻⁸ F,

Ro=2Ω,

r_(Mg) =300Ω.

As seen from the figure, the value Y increases as the pulse width T_(ON)increases; provided that the frequency f of the inverter is about 100kHz and the operating range of the pulse width T_(ON) is approximatelyfrom 4 to 5 microseconds, the value Y is approximately in linearrelationship with the pulse width T_(ON). Thus, under these conditions,the increase in the output power P_(OUT) given by equation (1) above isapproximately proportional to the increase in the pulse width T_(ON). Onthe other hand, the mean voltage signal Vf', which is obtained from thevoltage Vf corresponding to the magnetron current i_(Mg) by rectifyingand smoothing it by the rectifier 3a and the smoothing filter 3b asshown in FIG. 3b, is proportional to the magnetron output power P_(OUT).Thus, when the magnetron output power P_(OUT) decreases, the errorsignal Ve, the increase of which corresponds to the decrease in themagnetron output power P_(OUT), increases, because the decrease in theoutput power P_(OUT) increases, the mean voltage signal Vf' increases,thereby decreasing the error signal Ve. Thus, the pulse with T_(ON) alsodecreases to decrease the output power P_(OUT). Therefore, the magnetronoutput power P_(OUT) is maintained at a constant level determined by thesetting of the variable resistor 3c.

Further, the peak or maximum value i_(Mg) max during the stableoperation of the magnetron 1 is given, when ωT_(ON) >Z, by: ##EQU5##

FIG. 7 shows the relationship between the value ##EQU6## correspondingto the variable factors in the expression (2) and (2)' and the pulsewidth T_(ON), in the case where

n=10,

C=0.47×10⁻⁸ F,

Ro=2Ω,

r_(Mg) =300 Ω.

As seen from the figure, the value X is proportional to the pulse widthT_(ON) when the inductance L of the reactor L is large enough; forexample, in the case where the frequency f of the inverter is around 100kHz and the pulse width T_(ON) is limited within the range from about 4to 5 microseconds, the magnetron peak current i_(Mg) max can berepresented by a linear equation if the value of L is selected at 8miceohenries at which the value of X is approximately proportional tothe pulse width T_(ON) ; namely, i_(Mg) max is approximated by:

    i.sub.Mg max =K·(2Vo-V2/n)·T.sub.ON,     (3)

wherein K is the proportionality constant determined by the relationshipbetween X and T_(ON). The output voltage Vo of the filtering capacitorC1 appearing in the right hand side of expression (3) above is subjectto variation due to the variation in the AC voltage source E:

    Vo=V.sub.DC +ΔV,                                     (4)

wherein V_(DC) represents the pure DC, i.e., constant, component of thevoltage Vo and ΔV represents the AC component, i.e., variation, of thevoltage Vo. In order to maintain the peak current iMg max given by theapproximate equation (3) at a constant level irrespective of thevariation ΔV in the voltage Vo, T_(ON) should be varied to satisfy thefollowing equation:

    T.sub.ON =K.sub.1 /(2Vo-V.sub.Z /n)                        (5)

wherein K1 represents an arbitrary proportionality constant. Bysubstituting the right hand side of equation (4) into the right handside of equation (5) and expanding the right hand side of the equation(5) into Taylor series, i.e., into an infinite sum of the powers of ΔV,wherein the infinitesimal terms of degrees equal to or greater than 2are neglected, the pulse width T_(ON) is approximately expressed asfollows:

    T.sub.ON =K2-K3·ΔV,                         (6)

wherein K2 and K3 are constants determined by the values of K1, Vo, andn. On the other hand, the modulating signal Vp outputted from thesubtractor 3f to the PWM modulator 3h is given by:

    Vp=Ve'-Vin B,

wherein Ve' is constant in a stable operation and Vin is proportional tothe voltage Vo=V_(DC) +ΔV. Thus, the pulse width T_(ON) of the signal Vwoutputted from the modulator 3h, or that of the gate signals outputtedfrom the driver 3i, can be expressed as follows:

    T.sub.ON =K4-K5ΔV,                                   (7)

wherein K4 is a constant determined by the magnitude of the amplifiederror signal Ve' and the constant voltage component V_(DC) of thevoltage Vo, and K5 is a constant determined by the voltage signal Vinand the amplifying factor B of the amplifier 3g. Therefore, by selectingthe values of the constants K4 and K5 in equation (7) in such a way thatthey agree with the values of the constants K2 and K3 in equation (6),respectively, the peak current i_(Mg) max of the magnetron 1 can bemaintained at a constant level irrespective of the variation ΔV in thesmoothed DC voltage Vo outputted from the filtering capacitor C1. Inthis manner, the magnetron peak current i_(Mg) max is held substantiallyconstant even when the AC line voltage source E fluctuates. In otherwords, the inverter current flowing through the MOSFETs Q1 through Q4 isstabilized, thereby eliminating the danger of failures thereof.

Second and Third Mode: Simplified Inverter Switching Circuits

Referring now to FIGS. 8 and 9 of the drawings, a second and a thirdembodiment according to the present invention having a push-pull typeinverter switching circuit are described.

FIGS. 8 and 9 show a second and a third embodiment according to thepresent invention, respectively, both of which have a structure andoperation similar to that of the first embodiment, except for theinverter switching circuit and the position of the reactor. Thus, afull-wave diode bridge rectifier circuit 2 is coupled across thecommercial AC voltage source E, the output terminals of the rectifiercircuit 2 being coupled across the series connected resistors R1 and R2constituting a voltage devider and across the capacitor C1 constitutinga smoothing filter. The inverter switching circuit, however, consists ofa pair of MOSFETs Q1 and Q2, and diodes D1 and D2 coupled across thesource and the drain terminal thereof for reverse currents. In the caseof the second embodiment shown in FIG. 8, the source and the drainterminal of the MOSFETs Q1 and Q2 are coupled across the negativeterminal of the capacitor C1 and the terminals of the primary winding Pof the step-up transformer T, respectively, the positive output terminalof the capacitor C1 being coupled to the center tap 0 of the primarywinding P of the transformer T. Thus, in this second embodiment, thereactor L having a function corresponding to that of the reactor L ofthe first embodiment is inserted in series with the secondary winding Sof the transformer T, the capacitor C2 and the diode D3 being coupled inseries with the secondary winding S and the reactor L to form arectifier circuit corresponding to the rectifier circuit consisting ofthe capacitor C2 and the diode D5, as in the case of the firstembodiment. In the case of the third embodiment shown in FIG. 9, theprimary winding of the transformer T is devided into two portions P1 andP2; a mutual inductance M having a pair of magnetically coupled coils M1and M2 is coupled across the terminals 01 and 02 without dot marks inthe figure, the mutual inductance M effecting a function correspondingto that of the reactor L of the first embodiment. Thus, the MOSFETs Q1and Q2 are coupled across the negative terminal of the capacitor C1 andthe dotted terminals 03 and 04 of the windings P1 and P2, respectively;the positive terminal of the capacitor C1 is coupled to the terminalbetween the two coils M1 and M2 of the mutual inductance M. The circuitcoupled to the secondary winding S of this third embodiment is similarto that of the first embodiment.

In both second and third embodiment, the voltage devider consisting ofthe series connected resistors R1 and R2 outputs a voltage Vincorresponding to the output voltage Vo of the capacitor C1 to the PWMcontrol circuit 3; the current detector 4 detects the current flowingthrough the secondary winding S of the transformer T and output avoltage Vf corresponding thereto to the control circuit 3. The controlcircuit 3, which has a structure and an operation similar to those ofthe control circuit 3 of the first embodiment, outputs gate signalsalternately to the MOSFETs Q1 and Q2, and alternately turns them on andoff, modulating the pulse width thereof. Thus, in the positivehalf-cycle in which the MOSFET Q1 is turned on and the MOSFET Q2 isturned off, the induced voltage in the secondary winding S of thetransformer T has a polarity agreeing with that of the diode D3;consequently, the induced current in the secondary winding S charges thecapacitor C2 during the positive half-cycle. In the negative half-cycle,the MOSFET Q2 is turned on, while the MOSFET Q1 is turned off; thus, thepolarity of the induced voltage in the secondary winding S is reversed,and is applied across the magnetron 1 together with the voltagedeveloped across the capacitor C2. The resulting voltage V2 causing thecurrent i_(Mg) to flow from the anode An to the cathode K of theMagnetron 1.

Fourth Mode: Preferred Inverter Frequency

Referring now to FIG. 10 of the drawings, a fourth embodiment accordingto the present invention is described.

The power supply circuit shown in FIG. 10 has a structure similar tothat of the second embodiment. Thus, the input terminals of the diodebridge full-wave rectifier circuit 2 are coupled across the outputterminals of the commercial AC voltage source E; the output terminals ofthe rectifier circuit 2 are coupled across the capacitor C1 constitutingthe smoothing filter circuit. The inverter switching circuit 5 comprisesa pair of MOSFETs Q1 and Q2 and diodes D1 and D2 coupled thereacross inreversed polarity. The MOSFETs Q1 and Q2 are coupled across the negativeterminal of the capacitor C1 and the terminals 01 and 02 of the primarywinding P of the step-up transformer T; the positive terminals of thecapacitor C1 is coupled to the center tap 0 of the primary winding P ofthe transformer T. The voltage doubler half-wave rectifier circuitconsisting of a capacitor C2 and a diode D3 connected in series iscoupled across the secondary windings S of the transformer T, to supplypulsing DC voltage V2 to the magnetron 1 provided with a cathode K andan anode An. The filament voltage source 1a for the magnetron 1 isexplicity shown in FIG. 10.

However, the fourth embodiment is simplified compared with the second orthe third embodiment in certain respects. Namely, no reactor L or mutualinductance M is provided in the circuit. Further, no current detector isprovided for detecting the current flowing through the secondary windingS of the transformer T, and the voltage Vo developed across thecapacitor C1 is directly supplied to the control circuit 30 and thedriver circuit 31.

The control circuit 30 and the driver circuit 31 together correspond tothe control circuit 3 of the first through the third embodiment. Thecontrol circuit 30 may primarily be constituted by TL-494, an IC forswitching regulator source, produced by TI company, for example, andoutputs Vw1 and Vw2 alternately to the driver circuit 31; the pulsewidth of these pulses Vw1 and Vw2 can be varied in response to thevoltage Vo supplied thereto. The driver circuit 31 outputs gate signalsalternately to the MOSFETs Q1 and Q2 in response to the pulses Vw1 andVw2 to turn them alternately on and off.

Thus, current alternately flows through the upper and the lower half ofthe primary winding P from the center tap 0. Consequently, an AC voltageis induced in the secondary winding S of the transformer T which isstepped up by a factor equal to the ratio of the number of turns of thesecondary winding S to the number of turns of the primary winding Pbetween the center tap 0 and the terminal 01 or 02 of the transformer T.This AC voltage induced in the secondary winding S is converted into aunidirectional pulsing current by the voltage doubler half-waverectifier circuit consisting of the capacitor C2 and the diode D3, andis applied therefrom across the magnetron 1; thus, the magnetron isdriven by a pulsating current. Consequently, the microwave generated bythe magnetron 1 pulsates. FIG. 11 shows the change of the output powerP_(OUT) of the microwave generated to time plotted along the abscissa.

The reason why the output power P_(OUT) of the magnetron 1 takes thewaveform as shown in FIG. 11 is as follows. In the half-cycle of theswitching circuit 5 in which the MOSFET Q2 is turned on, the inducedvoltage in the secondary winding S has a polarity which agrees with theforward direction of the diode D3. Thus, in this half-cycle, thecapacitor C2 is charged by the induced current flowing through the diodeD3 and the secondary winding S; no voltage is applied across themagnetron 1. In the succeeding half-cycle in which the MOSFET Q1 isturned on while the MOSFET Q2 is turned off, a voltage having a reversedpolarity with respect to the diode D3 is induced in the secondarywinding S of the transformer T. Thus, the diode D3 is turned off, andthe sum of the voltages induced in the secondary winding S and developedacross the capacitor C2, which is charged in the previous half-cycle, isapplied across the magnetron 1. In FIG. 11, t1 corresponds to the timein which the MOSFET Q1 is turned on, to drive the magnetron 1 by the sumof the induced voltage in the secondary winding S and the voltagedeveloped across the capacitor C2; t2 represents the time in which theMOSFET Q1 is turned off. Thus, the waveform of the microwave outputpower of the magnetron 1 consists of a train of pulses having a pulsewidth t1 and recuring at the period To=t1+t2, as shown in FIG. 11.

The magnetron 1 is disposed in a microwave discharge light sourceapparatus, such as those shown in FIGS. 1a and 1b, which comprise aspherical electrodeless bulb. Then, the inverter switching frequency f,i.e. the frequency f=1/To of the pulses of the microwave output powerP_(OUT) of the magnetron 1 expressed in kHz, is preferred to be not lessthan the magnitude 1500/D; namely; it is preferred that

    f>1500/D,                                                  (8)

wherein

D=(the diameter of the electrodeless bulb expressed in millimeters).

The reason therefor is as follows.

An experiment has been conducted utilizing a microwave discharge lightsource apparatus shown in FIG. 1a, wherein the bulb 4 has a diameter of30 mm, 100 mg of mercury being encapsulated therein as an light emittingsubstance. When the magnetron input power is set at 1.5 kW and theinverter switching frequency f is varied in the range of from about 10to 20 kHz, the discharge in the bulb become unstable in intervals ofsubstantial widths within this frequency range.

This unstability in the discharge is inferred to be due to an acousticresonance phenomenon similar to that caused by sound waves in the bulbhaving electrodes, which is clarified in Shomeigakkaishi (IlluminationSociety Review) vol. 67 No. 2, pp. 55 through 61. However, in the caseof a discharge bulb having electrodes, the discharge therein is an arcdischarge caused across the two electrodes, the discharging regiongenerally forming a line across the electrodes. In contrast thereto, thebulb which is utilized in the light source apparatus according to thepresent invention is electrodeless; the discharge therein is maintainedby the microwave energy entering thereinto through the wall thereof:when the bulb has a spherical shape as in the apparatus of FIG. 1a, thedischarge therein is also spherical. Thus, the state of the dischargecaused in the electrodeless bulb by a microwave according to the presentinvention is completely different from that of the discharge bulb havingelectrodes; consequently, the acoustic resonance phenomenon of theelectrodeless bulb must also differ from that of the bulb havingelectrodes. More explicitly, it is known that the acoustic resonancephenomenon depends on the velocity of the sound wave in the dischargemedium gas and on the dimension and the shape of the discharge bulb; thevelocity of the sound wave varies with the temperature and the pressureof the gas through which it is propagated. Thus, as described above, dueto the difference in the states of the discharge in the electrodelessbulb and the bulb with electrodes, the temperatures and the temperaturedistributions of the gas, or the distributions of the velocity of thesound waves in these two types of bulbs, are different from each other.

In spite of these differences, certain conclusions may be drawn from theexperiments conducted by the inventors. Namely, in an experimentutilizing the apparatus of FIG. 1a having a spherical electrodeless bulb30 mm across (D=30 mm), wherein the inverter switching frequency f wasvaried to test the stability of the discharge in the bulb in varyingfrequency, it has been observed as follows: when the frequency f is lessthan or equal to 50 kHz, the intervals of frequency f in which thedischarge is unstable occupy considerable proportions; when thefrequency f is greater than 50 kHz, however, the widths of theseintervals shrinks rapidly as the frequency f is increased. Thus, underthe above condition, it can be concluded that the stable discharge canbe maintained in the electrodeless bulb if the discharge in the bulb iscaused by the microwave generated by a magnetron driven at a switchingfrequency not less than 50 kHz. From this particular example, generalformula for the preferred value of the inverter switching frequency fcan be obtained. Namely, the frequency f at which an acoustic resonancephenomenon takes place is proportional to the sound wave velocity C inthe discharging gas and inversely proportional to the diameter D of thedischarge bulb:

    f∝C/D.

The sound wave velocity C in the gas, however, varies little where themercury in the electrodeless bulb attains a relatively high pressure,i.e. 1 atmosphere, in operation. Thus, the resonating frequency isinversely proportional to the diameter D of the bulb. In the aboveexperiment, it has been decided that the resonance is substantiallyreduced when the frequency f is not less than 50 kHz at D=30 mm. Thus,it can be generally concluded that the acoustic resonance causingunstability in the discharge can be substantially reduced if thefrequency f satisfies the following inequality:

    f (kHz)≧1500/D,                                     (8)

wherein D represents the inner diameter of the bulb in millimeters.

Further, if the frequency f satisfies equality (8) above, there is nodanger that the discharge in the bulb is extinguished in the timeintervals t2 between the pulses of the microwave output power shown inFIG. 11, as explained in what follows:

In the power supply circuit of FIG. 10, a half-wave voltage doublerrectifier circuit consisting of a capacitor C2 and a diode D3 is used torectify the voltage induced in the secondary winding S of thetransformer T. Thus, as shown in FIG. 11, the microwave output powerP_(OUT) is reduced to zero in the time intervals t2 between the timeintervals t1 in which the MOSFET Q1 is turned on. The duration of thetime intervals t2, however, does not exceed 1 millisecond, provided thatthe frequency f is not less than 1 kHz, even if the pulse width t1 isdecreased in PWM control thereof. On the other hand, the so-calledafter-glow of the discharge, during which the discharge is maintainedafter the energy supply thereto ceases, is not less than about 1millisecond, provided that the plasma generating medium in the bulbconsists of substances usually utilized in a discharge bulb, i.e., arare gas, or a combination of rare gas and mercury or other metal. Thus,if the length of the time intervals t2 in which no microwave energy issupplied to the bulb does not exceed 1 millisecond, the discharge in thebulb is maintaining through the time interval t2 because, after thesupply of the microwave energy carried by a pulse thereof ceases, thedischarge in the bulb is maintained by the after-glow until thesucceeding pulse of microwave energy is supplied thereto. By the way, ifthe frequency f satisfies inequality (8) above, the diameter D of thebulb must be as great as 1500 mm to reduce the frequency f to 1 kHz atwhich the length of the time intervals t2 can not exceed 1 milliseconds.However, the diameter D of the bulb does not exceed 100 mm in practicalelectrodeless discharge light source apparatus. Thus, if the frequency fsatisfies inequality (8), the length of time intervals t2 during whichthe microwave energy supply ceases does not exceed 1 millisecond in apractical electrodeless discharge bulb; consequently, there is no dangerthat the discharge is extinguished between the microwave energy supplypulses.

Fifth Mode: Preferred Ratio of the Peak to the Mean Magnetron Current

Referring now to FIG. 12 of the drawings, a fifth embodiment accordingto the present invention is described.

The fifth embodiment shown in FIG. 12 has a structure and an operationsimilar to those of the first embodiment shown in FIGS. 3a and 3b. Thus,the input terminals of a diode bridge full-wave rectifier circuit 2consisting of four diodes Do connected in bridge circuit are coupledacross a commercial AC voltage source E; a smoothing filter circuit 3consisting of a choke coil Lo and a smoothing capacitor Co connected inseries is coupled across the output terminals of the rectifier circuit2. The output terminals of the filter circuit 3 are coupled to the inputterminals of the inverter switching circuit 4 comprising four MOSFETs Q1through Q4 connected in bridge circuit relationship; the switchingcircuit 4 further comprises four diodes D1 through D4 coupled across thesource and the drain of the MOSFETs Q1 through Q4 to allow currents inreverse direction, respectively, and a series connection of a capacitorand a resistors C1 and R1 through C4 and R4 coupled across each one ofthe MOSFETs Q1 through Q4, in parallel with the diodes D1 through D4,respectively. The output terminals of the switching circuit 4 arecoupled across the primary winding P of the step-up transformer T.Further, a half-wave rectifier circuit 5 consisting of a capacitor C5and a diode D5 connected in series is coupled across the secondarywinding S of the transformer T; a capacitor-diode circuit 6 is coupledacross the diode D5 of the rectifier circuit to reduce high frequencycomponents of the output of the rectifier circuit 5, the capacitor-diodecircuit 6 consisting of a capacitor C6 and a diode D6 connected inseries. The diode D6 has a forward direction that agrees with thedirection of the magnetron current i_(Mg) and supresses the current inreverse direction therethrough; the capacitor C6 is coupled across thecathode K and the anode An of the magnetron 1 to reduce high frequencycomponents of the current flowing through the magnetron 1. The magnetron1 is provided with a filament (or heater) voltage supply lines h havingnoise-filtering capacitors Cf and inductors Lf.

The current detector 7 inserted between the anode An of the magnetron 1and the positive terminal of the capacitor C6 detects the current i_(Mg)flowing through the magnetron 1, and outputs a voltage Vf correspondingthereto to the control circuit 8. The control circuit 8 has a structuresimilar to that of the control circuit 3 of the first embodiment shownin FIG. 3b, and outputs gate signals Vg1 through Vg4 to the gateterminals g1 through g4 of the MOSFETs Q1 through Q4, respectively, ofthe inverter switching circuit 4, through an operation interruptioncircuit 9. The operation interruption circuit 9 comprises: a diodebridge full-wave rectifier circuit 9a having input terminals coupledacross the AC voltage source E, a Zener diode Zn coupled across theoutput terminals of the rectifier circuit 9a through a resistor R; fourseries-connected diodes D7 through D10 in parallel circuit with theZener Zn; and four transistors T1 through T4. Thus, the operationinterruption circuit 9 detects the zero phases of the commercial ACvoltage source E, and suppress the gate signals Vg1 through Vg4 in theneighborhoods of the zero phases of the AC voltage E to interrupt theswitching operation of the inverter switching circuit 4 in the same timeintervals; thus, the circuit 9 excepts the neighborhoods of the zerophases of the AC voltage E as the operation interrupting periods of themagnetron 1.

The operation of this fifth embodiment shown in FIG. 12 is as follows.

When the rectifier circuit 2 is electrically coupled to the voltagesource E through a switch, etc., the AC voltage E is rectified by therectifier circuit 2 into a pulsating DC voltage; this pulsating DCvoltage outputted by rectifier circuit 2 is smoothed into asubstantially constant voltage by the filter circuit 3 and outputtedtherefrom to the switching circuit 4. The control circuit 8 alternatelyoutputs gate pulse signals Vg1 and Vg4 and gate pulse signals Vg2 andVg3 at a predetermined frequency, e.g., at 100 kHz, the pulse width ofthese gate signals Vg1 through Vg4 being modulated to maintain theoutput power of the magnetron 1 at a predetermined level. Thus, theMOSFETs Q1 and Q4 and the MOSFETs Q2 and Q3 are alternately turned onand off; as a result, the current i₁ flowing through the primary windingP of the transformer T changes its direction at the switching frequencyof the MOSFETs Q1 through Q4, thereby inducing a square waveform ACvoltage of the same frequency in the secondary winding S of thetransformer T. The voltage doubler half-wave rectifier circuit 5 coupledacross the secondary winding S outputs a pulse-shaped voltage in eachhalf-cycle of the switching circuit 4 in which the MOSFETs Q1 and Q4 arereturned on, the magnitude of the voltage outputted by the rectifiercircuit 5 being substantially two times as great as the voltage inducedin the secondary winding S. This pulsating voltage outputted in saidhalf-cycles of the inverter switching circuit 4 by the rectifier circuit5 is applied across the capacitor C6 through the diode D6; when thisvoltage outputted from the rectifier circuit 5 charges the capacitor C6to the operating (or cut-off) voltage of the magnetron 1, the magnetrondriving current i_(Mg) begins to flow through the magnetron 1. Thus,microwave is generated by the magnetron 1, and is supplied to anelectrodeless bulb (not shown) to cause a discharge and luminescencetherein.

The operation interruption circuit 9, as described above, supresses thegate signals Vg1 through Vg4 during the operation interruption intervalsin the neighborhood of the zero phases of the AC voltage source E,typically at 50 to 60 Hz, and stops the operation of the magnetron 1 inthese operation interruption intervals. In this embodiment, the lengthof the operation interruption intervals is set at about 0.5milliseconds. The purpose of establishing these operation interruptionintervals of about 0.5 milliseconds in each half-cycle of the AC voltagesource E is as follows: the magnetron 1 may fall into an abnormaloperation, such as an abnormal oscillation; if this happens, themagnetron 1 does not recover the normal stable operation by itself;thus, it is desirable to establish certain time intervals in which theoperation of the magnetron 1 is stopped.

Referring now to FIG. 13, the operation of the circuit of FIG. 12 isexplained more explicity.

The gate signals Vg1 through Vg4 have waveforms as shown in FIG. 13 (a)and (b); the pulses Vg2 and Vg3 are outputted by the control circuit 8in the half-cycle Tp to turn on the MOSFETs Q2 and Q3; the pulses Vg1and Vg4 are outputted by the control circuit 8 in the half-cycle Tn toturn on the MOSFETs Q1 and Q4. The pulse width T_(ON) of these pulsesVg1 through Vg4 are modulated in PWM (pulse width modulation) control bythe control circuit 8 to maintain the mean output power of the magnetron1 substantially at a predetermined level. The frequency f of thesepulses Vg1 through Vg4, typically about 100 kHz, which is referred to asthe inverter switching frequency, is equal to the reciprocal 1/To of theperiod To of these pulse signals Vg1 through Vg4. When the inverterswitching frequency f is set at 100 kHz, the pulse width T_(ON) ismodulated in a range of from about 3 microseconds to about 4microseconds.

The operation of the circuit in the half-cycle Tp shown in FIG. 13 is asfollows. When the MOSFETs Q2 and Q3 are turned on by the pulses Vg2 andVg3 in the half-cycle Tp, the current i₁ in the primary winding P of thetransformer T flows in the direction opposite to that shown by the arrowin FIG. 12. Thus, the voltage Vs induced in the secondary winding S ofthe transformer T has a polarity shown by the arrow in FIG. 12. Theinduced voltage Vs rises rapidly substantially to the level n Vodetermined by the step-up, ratio n of the transformer T and the voltageVo supplied by the filter circuit 3, as shown in FIG. 13(d). The currenti_(S), however, rises gradually from substantial zero to a maximumduring the time T_(ON) in which the MOSFETs Q2 and Q3 are turned on,due, for example, to leakage inductance, i.e., self-inductances of theprimary and the secondary winding P and S, of the transformer T, asshown in FIG. 13(c). In the same time period T_(ON) in the half-cycleTp, this induced current i_(S) in the secondary winding S rapidlyreturns to substantial zero as shown in FIG. 13 (c). The voltage Vsacross the secondary winding S, however, is kept substantially at thelevel n Vo to which the capacitor C5 has been charged during the timeinterval T_(ON), as shown in FIG. 13 (d).

In the succeeding half-cycle Tn, the circuit of FIG. 12 operates asfollows. When the gate pulse signals Vg1 and Vg4 are outputted by thecontrol circuit 8, the MOSFETs Q1 and Q4 are turned on. Thus, thecurrent i₁ flows in the primary winding P in the direction shown by thearrow in FIG. 12; the polarities of the induced current i_(S) andvoltage Vs are reversed with respect to those of the precedinghalf-cycle Tp, as shown in FIGS. 13 (c) and (d). Thus, the outputvoltage of the rectifier circuit 5 rises to the sum of the inducedvoltage Vs in the secondary winding S and the voltage to which thecapacitor C5 thereof is charged in the preceding cycle Tp; this outputvoltage of the rectifier circuit 5 is applied across the capacitor C6,which is already charged in the polarity shown in FIG. 12 in precedinghalf-cycles Tn. Thus, the voltage V_(Mg) across the magnetron 1, whichis substantially equal to the voltage developed across the capacitor C6,has a waveform shown in a solid curve in FIG. 13 (e); the maximumvoltage level Vmax of the magnetron voltage V_(Mg) is attained near theend of the time period T_(ON). (The waveform of the magnetron voltageV_(Mg) in the conventional circuit according to FIG. 2b is shown in adotted curve therein for comparison's sake; the maximum voltage thereofis indicated by V'max.) When the magnetron voltage V_(Mg) rises abovethe operating or cut-off voltage Vz, the magnetron current i_(Mg) beginsto flow through the magnetron 1, and is maintained during the time inwhich the voltage V_(Mg) is above the operating voltage level Vz, asshown in a solid curve in FIG. 13(f). The mean magnetron current i_(o)shown therein substantially corresponds to the means output power Po ofthe magnetron output power P_(OUT), as the increase V=Vmax-Vz in themagnetron voltage V_(Mg) above operating voltage level Vz is smallcompared with the magnitude of the cut-off voltage Vz. The magnetroncurrent i_(Mg) attains its maximum i_(max) corresponding to the maximumvoltage Vmax of the magnetron voltage V_(Mg). (The dotted curve in FIG.13 (f) shows the magnetron current having the same mean value i_(o) inthe case of the conventional circuit according to FIG. 2b, the maximumvalue thereof being indicated by i'_(max).)

As shown in solid and dotted waveforms shown in FIGS. 13 (e) and (f),the maximum or peak values Vmax and i_(max) of the magnetron voltageV_(Mg) and the magnetron current i_(Mg) of the circuit of FIG. 12 isreduced compared with those V'max and i'_(max) of the conventionalcircuit according to FIG. 2b; this is primarily due to the presence ofthe capacitor C6. As the magnetron current waveforms shown in solid anddotted curves in FIG. 13 (f) both have the same mean value i_(o), theratio i_(max) /i_(o) of the peak to the mean value of the magnetroncurrent i_(Mg) in the circuit of FIG. 12 according to the presentinvention shown by the solid curve is equal to 2.8, while that of themagnetron current in the case of the conventional circuit of FIG. 2bshown by the dotted curve is equal to 4.2. Thus, in the circuit of FIG.12, the ratio i_(max) /i_(o) and, therefore, the high frequencycomponents of the magnetron current i_(Mg) are greatly reduced comparedwith those taking place in conventional power supply circuits for amagnetron.

FIG. 14 shows further illustrative examples showing the reduction of theratio of the peak to the mean value of the magnetron current in thecircuit of FIG. 12 according to the present invention. Namely, the solidand the dotted curves in FIGS. 14 (a) through (c) show the waveforms ofthe magnetron current having the same mean value i_(o) ; the cases ofthe circuit of FIG. 12 are shown in solid curves; those of theconventional circuit of FIG. 2b are shown in dotted curves. The curvesin FIG. 14 (a) correspond to the case where the commercial AC linevoltage E is 10% under the rate level; those in (b) to the case wherethe voltage E is at the rate level; those in (c) to the case where thevoltage E is 10% above the rate level. The pulse width T_(ON) has beenmodulated to keep the mean value of the magnetron currents i_(Mg) shownin FIGS. 14 (a) through (c) at the same level i_(o). The ratio i_(max)/i_(o) of the peak to the mean value of the magnetron current i_(Mg) inthe case of the embodiment according to the present invention shown insolid curves in FIG. 14 is equal to: 2.0 where the voltage E is 10%under the rated level, as shown in (a); 2.86 where the voltage E is atthe rated level, as shown in (b); 3.4 where the voltage E is 10% abovethe rated level, as shown in (c). On the other hand, the same ratioi_(max) /i_(o) in the case of the conventional circuit according to FIG.2b is equal to 2.6, 4.2, and 7.0, when the voltage E is 10% under, equalto, and 10% above the rated level, respectively, as shown in dottedcurves in FIGS. 14 (a) through (c), respectively.

When the ratio i_(max) /i_(o) of the peak to the mean magnetron currentbecomes greater than 3.75, namely, if

    i.sub.max /i.sub.o >3.75,                                  (9)

flickerings are observed in the discharge in the electrodeless dischargebulb which is caused by the microwave generated by such magnetroncurrent. Thus, in the case shown in FIG. 13 (f), the magnetron currentshown in solid curve according to the present invention causes noflickering in the discharge in the electrodeless bulb; the magnetroncurrent in the case of the conventional circuit shown in dotted curve,however, causes flickering in the discharge therein. Similarly, themagnetron currents shown in solid curves in FIGS. 14 (a) through (c)according to the present invention cause no flickering in the discharge;those in dotted curves of the conventional circuit shown in FIGS. 14 (a)through (c) all cause flickering; that shown in (c) causes intenseflickering in the discharge.

FIG. 15 shows a result of an experiment which shows the critical meaningof inequality (9) above. Namely the curve of FIG. 15 shows the changeobserved in the intensity of flickering in the arc of the discharge inthe electrodeless bulb, with respect to the peak to the mean magnetroncurrent ratio i_(max) /i_(o), plotted along the abscissa, wherein theinverter switching frequency f has been set at 100 kHz, and the meanmicrowave output power at 850 W in the circuit according to FIG. 12.From the experimental result shown in FIG. 15, it can be concluded thatno flickering occurs if the ratio i_(max) /i_(o) is not greater than3.75, namely, if

    i.sub.max /i.sub.o <3.75;                                  (10)

and that the intensity of flickering increases abruptly when the ratioi_(max) /i_(o) exceeds 3.75, the flickering becoming intense when theratio i_(max) /i_(o) reaches 4.2.

As described above, the existance of the capacitance of the capacitor C6in the circuit of FIG. 12 is effective to reduce this peak to mean ratioi_(max) /i_(o) of the magnetron current i_(Mg). FIG. 16 shows therelationships of the frequency f (plotted along the abscissa in kHz) andthe capacity of the capacitor C6 (plotted along the ordinate inmicrofarads) which is effective in supressing the occurrence offlickering in the discharge, i.e, in reducing the ratio i_(max) /i_(o)to a level satisfying inequality (10) above; the three curves correspondto the cases in which the mean magnetron output power Po is equal to 680W, 850 W, and 940 W, respectively. The results shown in FIG. 16 wereobtained by an experiment in which the circuit according to FIG. 12 wasused to supply microwave to a spherical electrodeless discharge bulb 30mm across, in which sodium iodide, mercury, and argon were encapsulated.

While description was made of particular embodiments according to thepresent invention, it will be understood that many modifications may bemade without departing from the spirit thereof; the appended claims arecontemplated to cover any such modifications which fall within the truespirit and scope of the present invention. For example, the inverterswitching circuit may be constituted by a half bridge circuit ormonolithic forward circuit instead of full bridge circuit or push-pullcircuit. Further, the switching circuit may comprise, instead of theMOSFETs utilized in the embodiments described above, power transistorsSIT or GTO, SI thyristors, or magnetic amplfiers. Further still, theinductance L in the first and the second embodiment may be constitutedby a leakage inductance of the step-up transformer, i.e., theself-inductances of the primary and the secondary winding thereof. Inthe case of the fifth embodiment, instead of the capacitor C6, aninductance may be inserted in series with the magnetron to suppress thehigh frequency components in the magnetron current; alternatively, acombination of an inductance and a capacitance may be used for the samepurpose.

We claim:
 1. A circuit system adapted to supply microwave energy to amicrowave discharge light source apparatus including an electrodelessdischarge bulb, comprising:first rectifier means, adapted to be coupledto an AC voltage source of a relatively low voltage and frequency, foroutputting a rectified voltage of a relatively low voltage; filtermeans, coupled to said first rectifier means, for smoothing saidrectified voltage outputted from said first rectifier means, and foroutputting a smoothed rectified voltage; inverter means, coupled to saidfilter means, for converting said smoothed rectified voltage outputtedfrom said filter means to an AC voltage of a relatively high frequencyhaving a waveform of alternating pulses; pulse width modulation controlmeans for modulating a pulse width of said pulses of said AC voltageoutputted from said inverter means; a step-up transformer having aprimary winding coupled to an output of said inverter means, a secondarywinding of the step-up transformer outputting an AC voltage of saidrelative high frequency and of a relatively high voltage; secondrectifier means, coupled to said secondary winding of said step-uptransformer, for rectifying said AC voltage, of the relative highfrequency and the relative high voltage outputted from, said secondarywinding to a rectified voltage of a relatively high voltage, a magnetroncoupled to said second rectifier means, to be supplied with and operatedby said rectified voltage of the relative high voltage outputted fromsaid second rectifier means; and inductance means, operatively coupledto said step-up transformer, for supressing a rapid change in a level ofa current flowing through a winding of said step-up transformer.
 2. Acircuit system as claimed in claim 1, wherein said inverter meanscomprises a switching circuit including four transistors electricallyconnected in full bridge circuit relationship.
 3. A circuit system asclaimed in claim 1, wherein said inverter means comprises a switchingcircuit including a pair of transistors electrically connected inpush-pull circuit relationship.
 4. A circuit system as claimed in claims1 or 2, wherein said inductance means comprises an inductanceelectrically connected in series with said primary winding of saidstep-up transformer.
 5. A circuit system as claimed in claims 1 or 3,wherein said inductance means comprises an inductance electricallyconnected in series with said secondary winding of said step-uptransformer.
 6. A circuit system as claimed in any one of the claims 1through 3, wherein said inductance means comprises a leakage inductanceof said step-up transformer.
 7. A circuit system as claimed in claims 1or 3, wherein said primary winding of said step-up transformer comprisesa first and a second winding portion, and said inductance meanscomprises a mutual inductance electrically connected between said firstand second winding portion of said primary winding in series circuitrelationship.
 8. A circuit system as claimed in claim 1, wherein saidpulse width modulation control means comprises current detector meansfor detecting a current level of a current flowing through saidmagnetron, and means for varying said pulse width of said AC voltageoutputted from said inverter means in response to said current level ofthe current flowing through the magnetron detected by said detectormeans, thereby maintaining an output power of the magnetron at apredetermined level.
 9. A circuit system as claimed in claim 8, whereinsaid predetermined level is variable.
 10. A circuit system as claimed inclaim 1, wherein said first rectifier means comprises four diodeselectrically connected in bridge circuit relationship.
 11. A circuitsystem as claimed in claims 1 or 10, wherein said filter means comprisesa capacitor electrically connected across output terminals of said firstrectifier means.
 12. A circuit system as claimed in claim 1, whereinsaid second rectifier means comprises a diode and a capacitorelectrically connected in series coupled across terminals of saidsecondary winding of the step-up transformer.